Transistor switch device

ABSTRACT

A transistor switch device includes a main power transistor, an auxiliary solid-state switch, and an electricity replenishing means, wherein the electricity replenishing means is connected to the auxiliary solid-state switch which is connected between the collector and the base of the main power transistor so that the voltage drop at the collector of the main power transistor is reduced.

CROSS-REFERENCE TO RELATED INVENTIONS

This application is a continuation of Application Ser. No. 695,029,filed on June 11, 1976, now abandoned.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a transistor switch device capable ofoperation with a reduced voltage drop at the collector of the main powertransistor.

2. Description of the Prior Art

High-power transistors, in comparison with thyristors, have a selfturn-off function but cause a high collector voltage drop in thehigh-voltage element during conduction, resulting in a large loss. In aDarlington transistor circuit, the loss is especially large, amountingto two to three times that in a thyristor circuit. In applications wherelarge power is controlled or converted, the Darlington transistorcircuit has been considered impracticable in view of its need forconsiderable cooling and its low current utilizing efficiency.

With respect to FIGS. 1(a) and 1(b), there are shown circuit diagrams ofprior art power transistor switch devices. FIG. 1(a) shows aconventional Darlington circuit, comprising power sources 100a and 100b,a load 200, main power transistors 1a and 1b, and auxiliary transistors2a and 2b connected between the collectors and the bases, respectively,of the main power transistors 1a and 1b. In this Darlington circuit, thenecessary collector-emitter voltage V_(CE).sbsb.2 (ON) of the auxiliarytransistor 2 in the ON state is supplied from the collector-base voltageV_(CB).sbsb.1 (ON) of the main transistor 1 in the ON state and hencethe collector-emitter voltage V_(CE).sbsb.1 (ON) of the main transistor1 in the ON state becomes large causing the collector loss to beincreased during power supply. This has lowered the current utilizingefficiency and made cooling difficult. For these reasons, the Darlingtontransistor circuit as in FIG. 1(a) is not practical for use as alarge-capacity power switch device.

FIG. 1(b) shows another power transistor switch device in which thevoltage drop at the collector of the main transistor 1 in the ON stateas well as loss can be reduced. However, a rather large amount of poweris required from power sources B₁ and B₂ to drive the bases thereof. Thepower can be reduced but at the sacrifice of greater collector voltagedrop and loss in the main power transistor 1.

SUMMARY OF THE INVENTION

An object of the invention is to provide a transistor switch devicecomprising an electricity replenishing means connected to the auxiliarysolid-state switch in order to reduce the voltage drop in the maintransistor and to save the necessary capacity of the base drive powersource.

Another object of the invention is to provide a transistor switch devicein which the base drive power for the main transistor isself-sufficiently replenished.

The foregoing and other objects are attained in accordance with oneaspect of the present invention through the provision of a transistorswitch device comprising a main power transistor having a collector, abase and an emitter, an auxiliary solid-state switch, electricityreplenishing means, means connecting the auxiliary solid-state switchbetween the collector and the base of the main power transistor, meansconnecting the electricity replenishing means to the solid-state switchwhereby the voltage drop at the collector of the main power transistoris reduced.

BRIEF DESCRIPTION OF THE DRAWINGS

Various objects, features and attendant advantages of the presentinvention will be more fully appreciated as the same becomes betterunderstood from the following detailed description of the presentinvention when considered in connection with the accompanying drawings,in which:

FIGS. 1(a) and 1(b) are circuit diagrams showing prior art powertransistor circuits,

FIG. 2(a) is a circuit diagram showing one embodiment of the invention,and FIGS. 2(b) and 2(c) are diagrams of circuit components embodying theinvention,

FIG. 3(a) is an equivalent circuit diagram of FIG. 1(b) and FIG. 3(b) isan equivalent circuit diagram of FIG. 2(a),

FIGS. 4(a) through 4(c) are circuit diagrams showing electricityreplenishing means embodying the invention,

FIG. 5 is a circuit diagram of a switching device using a plurality ofpower transistors according to the invention,

FIG. 6 is a circuit diagram showing another embodiment of the invention,

FIGS. 7(a)-7(c) through 9 are diagrams showing operations of thecircuits of FIG. 6,

FIGS. 10(a) and 10(b) are circuit diagrams showing other embodiments ofthe invention,

FIGS. 11 and 12 are diagrams showing operations of the circuits of FIG.10,

FIGS. 13(a) and 13(b) are circuit diagrams showing another embodiment ofthe invention,

FIG. 14 is a circuit diagram showing another embodiment of theinvention; FIGS. 6 through 14(b) show examples of circuits operableindependently from DC power,

FIGS. 15(a) and 15(b) are circuit diagrams showing other embodiments ofthe invention,

FIG. 16 is a connection diagram showing one embodiment of the inventionusing a plurality of power transistors, and

FIGS. 17(a), 17(b), 18(a), 18(b) and 19 are circuit diagrams showingapplication examples using the principle of the invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Referring now to the drawings, wherein like reference numerals designateidentical or corresponding parts throughout the several views, and moreparticularly to FIG. 2(a) thereof, there is shown a transistor switchcircuit comprising main power transistors 1a and 1b, auxiliarysolid-state switches 2a and 2b connected between the collectors and thebases, respectively, of the main power transistors 1a and 1b, and areplenishing power source 40 connected in series to the auxiliary solidstate switches 2a and 2b. FIG. 2(a) shows an example wherein thereplenishing power source 40 is used in common for the two powertransistors. The auxiliary solid-state switches 2a and 2b may be oftransistor, gate-turn-off thyristor as in FIG. 2(b), thyristor (with anextinction circuit provided separately), or the like. The replenishingpower source 40 may only be capable of offering a voltage as low as 0.5to 2 V. According to the invention, even if the collector-base voltagesof the main transistors 1a and b are low, that is, the collector-emittervoltages thereof are low, the voltage drops in the auxiliary solid-stateswitches 2a and 2b (hereinafter briefly, auxiliary transistors) whilethey are in the ON state are replenished with the voltage from the powersource 40. As a result, the necessary collector potential of theauxiliary transistor 2 is duly maintained, and the current amplificationfactor hfe₂ remains sufficiently high. This serves to suppress thevoltage drop in the main transistor 1.

This operation will be described more specifically by referring to FIG.3. FIG. 3(a) is an equivalent circuit diagram of the prior art circuitas in FIG. 1(b). In FIG. 3(a), the reference Vd₃ denotes the minimumvoltage required of the base drive power sources 3a and 3b. The voltageVd₃ is the sum of the base-emitter voltage V_(BE).sbsb.1 of the maintransistor 1 and the collector-emitter voltage drop V₂ of the auxiliarytransistor 2, or Vd₃ =(V_(BE).sbsb.1 +V₂).

In the circuit of the invention as in FIG. 2(a), the voltage Vd₄ of thereplenishing power source 40 is lower than the collector potential(i.e., Vd₃) of the auxiliary transistor 2 by V_(CE).sbsb.1 which is thecollector-emitter voltage of the main transistor 1. That is, Vd₄ =(Vd₃-V_(CE).sbsb.1)=(V_(BE).sbsb.1 +V₂ -V_(CE).sbsb.1). In other words,under the application of the same base-drive voltage, the voltage V₂ ofthe auxiliary transistor 2 can be maintained high and its currentamplification factor can be increased, with the result that thecollector voltage drop V_(CE).sbsb.1 of the main transistor is reducedand the loss is reduced accordingly. Under the application of the samebase-drive current Ib₁ (the same base-drive power source current) andwith the same collector voltage drop V_(CE).sbsb.1 of the maintransistor, the necessary base-drive power source voltage Vd₄ is loweredby V_(CE).sbsb.1, permitting the necessary capacity of the base-drivepower source to be reduced. It is apparent that the collector voltagedrop V_(CE).sbsb.1 is lowered for the same base-drive power sourcecapacity. Thus, according to the invention, the voltage drop and thecollector loss in the main transistor are reduced to make cooling easy.For the same loss, greater current is allowed to flow in the circuit.Furthermore, because the current amplification factor hfe₂ of theauxiliary transistor can be maintained high, the external turn-on basecurrent Ib₂ may be made small.

Furthermore, as in FIG. 3, the main current I_(L) =Ic₁ +Ic₂. That is,the collector-emitter current Ic₂ is branched to the main transistors.In other words, the current Ic₂ passes through the main power path andsustains the power path current, enabling the current capacity of theauxiliary transistor to be effectively utilized in sharing the loadcurrent. The replenishing power source 40 of FIG. 2(a) may be of a DCsource such as a battery or other DC source provided by the use of arectifier or chopper. The replenishing power source used in the circuitas in FIG. 2(c) is the combination of a current source 41 and a diode42. (The diode 42 is used also as a branch means when the auxiliarytransistor switch is cut off.) The replenishing power source 40 and itscurrent source 41 will be described below in more detail.

FIGS. 4(a) to 4(c) show examples of replenishing power source 40.

In FIG. 4(a), the DC replenishing power source 40 having a droopingcharacteristic (current limiting characteristic) is obtained from an ACpower source 43 by way of an AC impedance 44 and a rectifier 45. Havinga current limiting characteristic, the DC replenishing power source 40operates half as the current source 41. When the impedance 44 is ofreactance X, the loss will be small.

In FIG. 4(b), the DC replenishing power sources 40a to 40n are obtainedfrom a high frequency AC power source 46 by way of a transformer 47 andrectifiers 45a to 45n. In this example, a current source 41 is obtainedby the use of a current limiting transformer such as a leakagetransformer in place of the transformer 47.

FIG. 4(c) shows an example wherein the replenishing power source(current source 41) 40 is obtained from an m-phase AC power source 43through rectifiers 45a to 45n by way of current transformers 48a to 48nand an impedance Z. The impedance Z may be of a power network from whichpower is supplied to the main power transistor circuits such as, forexample, the input circuit of the commercial AC power source used as theAC source 43 or may be provided separately.

Thus, according to the invention, the replenishing power is distributedby way of a transformer whereby the necessary capacity of the drivepower souce is reduced and the size of the transformer which occupies aconsiderable area of the base drive circuit is reduced.

The current source systems shown in FIGS. 2(b), 2(c) and 4 areadvantageous for the following reasons. When the replenishing sourcevoltage Vd₄ is too high and the impedance is low, the shunt current tothe side of the auxiliary solid-state switch is likely to become toolarge. However, a current source system having a drooping and currentlimiting characteristic does not give rise to such a problem. At thesame time, influence from variations in the voltage drop V₂ in theauxiliary solid-state switch is largely diminished. Furthermore, thecircuit of the invention operates at a minimum of drive voltage Vd₄ atall times for individual operating states, permitting the necessarypower to remain at a minimum. This makes it possible to reduce the sizeof the transformer to a minimum. Still further, the circuit as shown inFIG. 2(a) operates like a known Darlington circuit even without thereplenishing voltage Vd₄ if the period for which the replenishingvoltage is absent (ripple or dip) is short enough or the ratio of theperiod for which the replenishing voltage is absent or too low to theentire operating period is small enough. This means that the need forsmoothing the replenishing power source (base drive power source) isobviated or the smoother used is simplified. Also, for a shortovercurrent load, the collector voltage V_(CE).sbsb.1 of the main powertransistor rises to initiate the function of the self-sufficient powersupply. Hence, a shortage of the replenishing power does not immediatelycause an abrupt rise in the collector voltage of the main powertransistor and possible resultant breakdown.

FIG. 5 shows another embodiment of the invention wherein a plurality ofpairs of bidirectional conductive power control converter units (aplurality of pairs of half-bridge circuits) are used to realize a methodfor replenishing a plurality of power transistors with power. Thisembodiment can be applied to inverters, DC power control, time-ratiomodulation power amplifiers and the like.

In FIG. 5, the reference numeral 46 denotes a high frequency AC currentsource, and 48 a current transformer. Because the necessary power supplyvoltage may be low at a high frequency, the number of the secondarywindings of the transformer can be made as small as one turn(through-winding) to several turns. On the side of primary winding, itis desirable that the voltage be made high and the current be made lowbecause this approach facilitates forming the high frequency currentsource. (This principle does not apply to the instance in which a highfrequency current source is set up from a power source whose voltage isas low as ten or several volts.) For this reason, FIG. 5 shows anexample wherein the current transformer has more turns of winding on theprimary side than on the secondary side. In large power devices, powercan be supplied from a group of simple current transformers whosesecondary winding is of one to several turns, and hence powerdistribution through current transformers of a primary series connectiontype is desirable. In the embodiments as in FIGS. 4(c) and 5, thecurrent or voltage present due to turn-on at the main transistor or agroup of main transistors can be used as the input to the transformers47 or 48.

According to the invention, as described above, an auxiliary solid-stateswitch is installed between the collector and the base of the main powertransistor, and an electricity replenishing means is connected to theauxiliary solid-state switch whereby the collector voltage drop in themain power transistor is reduced. For the same collector voltage drop, alow power source voltage suffices for base drive, permitting thenecessary drive power to be reduced.

FIG. 6 is a circuit diagram showing another embodiment of the invention,which comprises a power circuit connecting a power source 100 (an ACpower source when one of the directions of an AC circuit is controlled)to a load 200, and a main power transistor 1 connected in series to atleast one power path X-Y of the power circuit which is controlled in theon-off mode. An auxiliary solid-state switch 2 is provided between thecollector and the base of the power transistor in order to supply basecurrent Ib₁. An electric transformer 3 is provided for derivingelectricity (such as voltage, current or power) produced in the powercircuit due to the function of the power transistor. The output ν₃ andI₂ of the transformer 3 is supplied to a branch circuit C₂ --2--B₁ ofthe auxiliary solid-state switch 2. This embodiment exemplifies a singleclosed-loop power circuit.

In this embodiment, the transformer 3 is a current transformer with itsprimary winding or primary through-conductor 31 inserted in the powerpath, and its secondary winding 32 (which includes the primary windingin the case of an autotransformer connection) inserted in series withthe solid-state switch branch circuit. The auxiliary solid-state switch2 is a transistor or thyristor. When the solid-state switch 2 is atransistor, this transistor switch can be used as an electricityreplenishing means (as will be described later), which makes it possibleto form a Darlington transistor circuit. In such an application,increase in the voltage drop in the main power transistor is obviated.In prior art techniques, the voltage drop in the main power transistoris great even in a two-stage (two subordinate collector common)Darlington circuit, and hence is not suited for large power capacityswitch devices. In a three-stage Darlington circuit, the voltage drop inthe main power transistor (with capacity greater than 50 A, 200 V) istoo high and the resultant loss is too large for practical use. Theauxiliary solid-state switch 2 is preferably a gate-turn-off typethyristor. When the ordinary thyristor is used for the auxiliarysolid-state switch 2, it is necessary to also provide an extinctionmeans.

In the embodiment using a current transformer as in FIG. 6, theauxiliary solid-state switch 2 is cut off. The shunt means for thesecondary current I₂ which flows when the solid-state switch 2 is cutoff is constituted of rectifier elements and resistors. The shunt means4 may be connected into the circuit as indicated by 4'. When the shuntmeans 4 is a rectifier element in the position indicated by the solidline, it is necessary to provide one or a plurality of such rectifierelements connected in series, depending upon the voltage drop in theauxiliary solid-state switch 2 in the ON state. Instead of thisrectifier element, a rectifier element whose forward voltage drop islarge, or a series of resistors (or nonlinear resistance elements) and arectifier element may be used. When the rectifier element is in position4' indicated by the dotted line, it is not necessary to provide aplurality of rectifier elements connected in series to increase thevoltage drop in the shunt means.

The rectifier elements 5a and 5b serve as base reverse biasing currentpaths for turn-off operation, through which the carrier stored in themain transistor 1 or auxiliary solid-state switch 2 is released. TheDarlington transistor switch 2 within the frame may be formed of aone-package transistor or a monolithic IC. The numeral 6 denotes anexternal turn-on control means including the reverse biasing path.

The core 33 of the current transformer 3 may be a ferrite system core orpermalloy system core for repetitive high frequency operation or asilicon steel system core for low frequency operation. In a currentpower network, an ordinary AC current transformer can be used as thetransformer 3. The core of the transformer 3 may be in the shape of cut,noncut, lamination, etc. with a multi-leg construction. In the circuitof FIG. 1, the transformer is inserted, together with the powertransistor 1, in series with the DC power path X-Y, to make thistransformer operate as a repetitive pulse current transformer as will bedescribed below.

In FIG. 7, assume that the initial state 1 where the main powertransistor 1 and the auxilairy solid-state switch 2 are ON is at thepoint 1 on the core B-H curve (represented by the exciting current I andthe number of interlinking fluxes φ) as shown in (a). This initial pointcorresponds to the timing 1 on the operating waveform shown in FIG. 8.

When a turn-on signal Ib₂ as in FIG. 8(a) is applied to the auxiliarysolid-state switch 2, the switch 2 turns on after a delay td₂. Theturn-on of the main power transistor 1, i.e., the presence of the fullload current I_(L) at the main power transistor 1, is delayed beyond thedelay td₂, to allow the voltage on the power path X-Y to be sustained.The turn-on delay time td₁ of the main power transistor 1 consists ofthe flux reset time t_(R) of the current transformer (or saturablereactor in a generic sense) and the delay td₁₀ after the increase ofbase current Ib₁ as in FIGS. 8(b) and 8(c).

A negative voltage VR is applied to the secondary winding 32 in FIG. 6(where the dot mark indicates the positive polarity) during the delayperiod td₁ (especially t_(R)), and the core flux falls along the B-Hloop as in FIGS. 7(a) (i) and 8(b) (i).

When the main transistor starts turning on before it reaches thenegative saturation level, the base current Ib₁ (which is equal to thecurrent I₂ of the auxiliary solid-state switch 2) of the main transistor1 increases along the solid line Q of FIG. 8(c) and thereafter steadycurrent transformer function starts. In this case, the minimum fluxlevel stands at point 2 of FIG. 7(a) or point 2 of FIG. 8(b).

When the main transistor turns on after it reaches the negativesaturation level, the base current Ib₁ of the main transistor 1 exhibitsa peak as indicated by the dotted line P in FIG. 8(c).

During the turn-on start delay period td₁ of the main transistor 1, theflux of the core 33 is reset and thereafter steady current transformerfunction starts.

In the event of a heavy load, i.e., a large I_(L), the voltage drop(substantial turn-on) of the main transistor 1 is delayed and theproduct of the negative voltage of the current transformer and the timefor which the negative voltage is sustained, that is, the flux resetvalue φ_(r) as in FIG. 7(b), increases. The greater the load I_(L), thelonger the delay td₁, with the result that the core reaches the negativesaturation level -φ_(s), causing a peak as indicated by the dotted lineP in FIG. 8(c).

The voltage drop delay time td₁ of the main transistor depends upon theratio of the excitation current I.sub.ε (indicated by the dottedencircled line in FIG. 8(c) to the secondary winding of the currenttransformer 3 (FIG. 6), to the power path current I_(L). The larger theratio I_(L) /I.sub.ε, the longer the delay td₁. Thus, the core flux isreset to the negative saturation level -φ_(s).

When the main transistor 1 turns on, the potential difference betweenthe power paths X-Y and C₁ -Y decreases whereby the winding 31 functionsas the primary winding, and the winding 32 as the secondary winding witha main current I_(L).

The potential V_(C2) at the terminal C₂ of the solid-state switch is thesum of the potential V_(B1) at the base B₁ of the main transistor 1 andthe voltage drop V.sub.(C.sbsb.2_(-E).sbsb.2.sub.) in the solid-stateswitch, and is larger than the collector voltageV.sub.(C.sbsb.1_(-E).sbsb.1.sub.) of the main transistor; that is,V.sub.(C.sbsb.1_(-E).sbsb.1.sub.) <V_(C2). Therefore, the currenttransformer voltage ν₃ is expressed as

    ν.sub.3 =V.sub.C2 -V.sub.(C.sbsb.1.sub.-E.sbsb.1.sub.) =V.sub.B1 +V.sub.(C.sbsb.2.sub.-E.sbsb.2.sub.) -V.sub.(C.sbsb.1.sub.-E.sbsb.1.sub.) (1)

In the prior art Darlington circuit, the transformer 3 is not used andν₃ =0. As a result, the condition V.sub.(C.sbsb.1_(-E).sbsb.1.sub.)=V_(B1) +V.sub.(C.sbsb.2_(-E).sbsb.2.sub.) serves as a limiting factor,causing the voltage drop V.sub.(C.sbsb.1_(-E).sbsb.1.sub.) in the maintransistor to be increased.

However, according to the invention, the transformer voltage ν₃ isreplenished by virtue of the current transformer 3. During thisreplenishing period ii in FIG. 7, the core flux rises along ii as inFIG. 7(a) and (b), where ν₃ =(ν₃₂ +ν₃₁). For this period, the flux φassumes a curve for the positive low voltage period ii as in FIG. 8(b).Then the solid-state switch current I₂ is: ##EQU1## where N₃₁ and N₃₂denote the numbers of turns of the current transformer windings. In FIG.6, the number of turns N₃₂ of the secondary winding includes those ofthe windings 31 and 32 since this transformer is an autotransformer.

As in FIG. 6,

    I.sub.L =I.sub.1 +I.sub.2                                  (3)

The base current Ib₁ of the main transistor 1 is supplied proportionalto the collector current I₁ of the main transistor. The ratio N₃₂ /N₃₁is determined as: ##EQU2## wherein hfe(min) denotes the minimum currentamplification factor applied for the maximum current in the maintransistor.

The time toneff of the steady period ii can be determined to besufficiently long relative to the flux reset time t_(R).

The flux level reaches the point 3 in FIGS. 2(a), (b) and 8(b)immediately before the main transistor 1 is turned off. When the reversebias -Ib₁ is supplied, the auxiliary solid-state switch 2 turns offafter its delay ts₂, causing the base current Ib₁ to be shunted to theshunt means 4. Then the main transistor 1 turns off also after its delayts₁. At this turn-off, the reverse bias pulse -Ib₁ is supplied from theturn-on control means 6 when necessary. In FIG. 8(a), the reference -Ib₂denotes a steady reverse bias current.

The core flux φ reaches the point 4 of FIG. 2(a) and (b) when theturn-off of the main transistor is completed. In this operation, theflux curve is nearly as 3 as in FIG. 7 when the shunt means 4 serves asthe rectifier element indicated by the solid line in FIG. 6. When theshunt means 4 is a resistor or rectifier element indicated by the dottedline, a high positive voltage is applied to the current transformer 3during the period ts₁ for which the auxiliary solid-state switch 2 is inthe OFF state and the main transistor 1 is in the ON state. As a result,the flux becomes more positive as 4 in FIG. 7(i c). If the flux level of3 is near the positive flux saturation level +φ_(s), the period ts₁ isreduced.

Then, for the period the main transistor 1 is OFF, the core flux changestoward the excitation current zero line (Y axis), at a velocitydepending upon the shunt means 4,4' or 4R. A series of the aboveoperations is repeated.

In the above operation, the condition ν₃ ≦0.5 to 1 V (where ν₃ is thereplenishing voltage) holds as long as the auxiliary solid-state switch2 is one transistor (not a Darlington connection) or one thyristor. Thisvalue of ν₃ is below the base-emitter voltage drop in the maintransistor. When the auxiliary solid-state switch is a Darlingtontransistor switch, ν₃ 1 to 2 V. As the voltage ν₃ becomes higher, thevoltage drop in the auxiliary solid-state switch is allowed to be higherand the voltage drop in the main transistor 1 can be lower. This meansthat the switch device of the invention is operable with a minimum ofpower loss, ease of cooling, and is applicable to large power capacitydevices.

As shown in FIG. 9(a), ON-OFF of the auxiliary solid-state switch isassumed to occur in the operation for the minimum OFF time toffmin. Theturn-off delay time ts of the main transistor 1 is included in thesubstantial ON time toneff. The flux reset time t_(R) of the currenttransformer and the turn-on delay time td₁₀ of the main transistor areincluded in the substantial OFF time toffeff. These time relations areshown in FIG. 9(b).

The current transformer operating condition dependent on the worst fluxreset is expressed as: ##EQU3## where E stands for the voltageimmediately before turn-on of the power path X-Y, and the flux resetvoltage is assumed to be VR=E.

Here the number of turns of the secondary winding is equal to thatbetween terminals C₁ and C₂, as in Eq. (2).

Also the following equation is established. ##EQU4##

That is, the turn-on duration ton is E/ν₃ times the flux reset timet_(R). The turn-on duration multiplying factor K can range from 100 to600 at a power path voltage of 200 to 300 V. Accordingly, when td₁₀≦t_(R), the maximum turn-on time ratio α max is: ##EQU5## This maximumvalue is readily feasible.

In the embodiment shown in FIG. 6, as described above, the switch devicefunctions substantially as a DC current transformer in spite of its useof a magnetic current transformer whereby a high turn-on ratio α ismaintained. In other words, the device makes DC power control possible.Even for low-frequency AC control, the use of a high-frequency(time-ratio control frequency) current transformer suffices. Hence thesize of the transformer can be reduced. For example, the use of aone-turn through type current transformer is sufficient, the size andconstruction of which may be about the same as a saturable reactor whichhas hitherto been utilized in suppressing the turn-on di/dt of thethyristor.

Furthermore, a considerable amount of base-drive power for the mainpower transistor is supplied self-sufficiently from the main power path.At the same time, the device of the invention offers the effect ofreducing the loss in the main power transistor. These advantages enhancethe usefulness of the device of the invention.

FIGS. 10(a) and 10(b) are circuit diagrams showing other embodiments ofthe invention, wherein a shunt means 7 is provided for shunting theexcitation current I.sub.ε to the base of the main transistor 1 for theflux reset period t_(R) at the beginning of turn on. This shunt means(transistor) is self-controlled through a tertiary winding 34 whichdetects the core flux reset voltage. Operating waveforms of the circuitsare shown in FIG. 11.

In FIGS. 10(a) and 10(b), when the auxiliary solid-state switch 2 startsturning on, the excitation shunt transistor 7 turns on by the emf of thetertiary winding 34 through an impedance 8, thereby maintaining the maintransistor 1 in the OFF state. (The shunt transistor 7 turns on beforethe main transistor turns on with the turn-on delay td₁.) As a result,the excitation current I.sub.ε of the current transformer 3 flows in theshunt transistor 7. The waveforms thereof are indicated by I₇ andI.sub.ε in FIG. 11(c).

When the core flux approaches near or reaches the negative saturationlevel, the tertiary winding voltage decreases to cause the shunttransistor 7 to turn off. Consequently, the current I₂ of the auxiliarysolid-state switch 2 is supplied to the base of the main transistor 1whereby the main transistor is turned on, as in FIG. 11(d) and (e). Whenthe main transistor turns on, the tertiary winding is at a slightlynegative voltage or a very low voltage and hence it is unlikely for theshunt means 7 to be turned on. The circuit will thereafter operate inthe same manner as the one shown in FIG. 6.

FIG. 10(b) shows an example wherein a voltage transformer 3 is installedin a turn-on power path beside the main transistor power path (maintransistor branch) X-Y.

In the embodiments shown in FIG. 10, the core flux can securely be resetto about the negative saturation level irrespective of the turn-on delayof the main transistor and irrespective of the ratio of the main currentI_(L) to the core flux current I.sub.ε.

FIG. 12 is a diagram showing operation for maximum continuous turn-on inthe circuit as in FIG. 10. In FIG. 12(b), the auxiliary solid-stateswitch 2 is kept turned on continuously. At the same time, theexcitation shunt means 7 is turned on for a given time t₇ at the maximumON time tonmax. The period t₇ is approximately equal to the turn-ondelay td₁₀ of the main transistor subtracted from the sum of theturn-off delay ts₁ of the main transistor and the flux reset time t_(R),or t₇ =ts₁ +t_(R) -td₁₀. Thus, as in FIG. 12(c), the main transistorrepeats ON-OFF at an ON-OFF ratio at which the main transistor can beconsidered to be substantially in a continued ON state.

In the circuit of FIG. 10, the excitation shunt means 7 can be turnedon-off by an external pulse of given width which synchronizes with theturn-on signal Ib₂. In this case, the need for the tertiary winding 34is obviated.

FIG. 13 shows another embodiment of the invention in connection withimprovements in the turn-on initial flux reset method. This circuitcomprises a diode or breakover switch 13, which may be a thyristor withanode igniting means, 5-layer semiconductor switch such as SSS,dynistor, PNPN switch, or the like. When the breakover voltage of thebreakover switch is higher than the base-emitter reverse peak voltage ofthe main transistor 1 and hence the voltage (during flux resetting) ofthe tertiary winding 34 is to be made high, a protective means such asreverse peak protection rectifier element 14, resistor 15, etc. shouldbe provided.

In FIG. 13(a), when the auxiliary solid-state switch 2 is turned on, anegative voltage is applied to the current transformer 3 before the maintransistor 1 starts turning on, causing a negative voltage to be inducedin the tertiary winding 34 whereby the breakover switch 13 turns on andthe auxiliary solid-state switch current I₂, i.e., the excitationcurrent I.sub.ε, flows in the excitation shunt means 7 which comprisesthe tertiary winding 34 and the breakover switch 13. During thisoperation, the base of the main transistor 1 is negatively biased.

When the voltage (negative) of the tertiary winding becomes small at theend of core flux reset, the base potential of the main transistor 1rises to cause the main transistor to turn on. Thereafter the circuitwill operate in the same manner as in FIG. 6. When the breakover switch13 is an element whose forward voltage drop and holding current IH arelarge and is capable of being readily turned off (such as having a V-Icharacteristic as indicated by (v) in FIG. 13(b)), the breakover switchturns off while the main transistor 1 is being turned on. In thismanner, this circuit operates as in FIG. 11.

FIG. 14 shows another embodiment of the invention wherein the maintransistor 1 is kept turned on continuously and the function of thecurrent transformer 3 is maintained continuously.

When the auxiliary solid-state switch 2 and the main transistor 1 arecontinuously in the ON state as in FIG. 14(b), an intermittent fluxreset switch 17 is turned on for the period t_(R) at intervals of timeT. In this operation, the period t_(R) can be controlled automaticallyunder self-control by the tertiary winding 34. Instead of the tertiarywinding 34, an external signal may be used to turn on the switch 17 fora given period t_(R). With self-control, it is desirable that theimpedance 18 be a differential element comprising, for example, aparallel resistor-capacitor circuit. During the period t_(R) for whichthe flux reset switch 17 is in the ON state, a negative voltage VR isapplied to the secondary winding voltage ν₃₂. During this operation, thediode 16 blocks the voltage VR, thereby preventing it fromshort-circuiting over the side of the main transistor. In this state,the secondary winding current I₂ is larger than the magneto-motive force(N₃₁.I₁) of the primary winding current I₁ by the excitation current,that is, ##EQU6## (where I.sub.ε is the excitation current changing fromnegative to positive). In other words, the function of the currenttransformer is maintained in the above state. When the flux reset switch17 is turned off, the circuit operation shifts to the foregoing steadyoperation. Thus the secondary winding current I₂, i.e., the base currentof the main transistor 1, assumes the waveform I₂ shown in FIG. 14(b),and the current transformer voltage ν₃₂ assumes the waveform ν₃₂.

As described above, the function of the current transformer 3 issteadily maintained while the main transistor is kept turned oncontinuously. In other words, the invention achieves its aim regarding"perfect turn-on by DC" or makes it possible to realize a 100% turn-onratio.

It is apparent that the switch circuit of the invention can beeffectively utilized within a finite turn-on time. The embodiments shownin FIGS. 1, 10, 13 and 14 are uniquely advantageous in that a means forapplying the pulse reset voltage VR is provided and the currenttransformer function is maintained by a single power transistor circuitby the use of a single core having substantially a full time range.Prior art techniques have failed in offering power transistor switchdevices capable of DC operation by the use of a transistor or the like;it has been thought theoretically impracticable. One example knownregarding such operation depends upon the use of plural-core AC(alternating) type construction.

FIGS. 15(a) and 15(b) are diagrams showing other embodiments of theinvention wherein the reference 3ν denotes a voltage transformer, 31νand 32ν the primary and secondary windings respectively, 36ν a resetwinding, and 201 a main current commutating diode (such as a flywheeldiode or a feedback rectifier) used while the main transistor is in theOFF state.

In FIG. 15, when the auxiliary solid-state switch 2 is turned on, themain transistor 1 turns on, a voltage is applied to the transformer 3ν,a voltage ν₃ (whose positive polarity is indicated by the dot (.)) isinduced across the secondary winding 32ν, and this voltage is suppliedto the auxiliary solid-state switch branch. As a result, the necessaryvoltage drop in the auxiliary solid-state switch and the base-emittervoltage drop in the main transistor are duly maintained even if thecollector voltage drop in the main transistor is insufficient, as in theembodiment shown in FIG. 1. In other words, the collector-emittervoltage drop and consequent power loss in the main transistor arereduced.

When the auxiliary solid-state switch 2 and the main transistor 1 areturned off, a reverse magneto-motive force is developed through thereset winding 36ν (used in common with the winding 32ν in FIG. 15(a)) bythe current of the main current path, such as by the commutating diode201, and thus the core flux is reset.

In this voltage transformer system, core flux reset cannot besufficiently done to make it impossible to operate the circuit over awide range of turn-on time ratios if the single DC circuit as in FIG. 15is employed. To solve this problem, it is necessary to use a means (suchas a power source) for providing a potential difference between anodesx₁ and x₂. This corresponds to an AC control as will be described later.Also, a means is provided for allowing a given voltage to be applied toor induced across the primary winding 31ν.

FIG. 16 is a circuit diagram showing another embodiment of the inventioncomprising a plurality of main power transistors. This circuit consistsessentially of a power network group A and a power network group B,which have a main transistor group 1A comprising mA numbers of maintransistors, and a main transistor group 1B comprising mB numbers ofmain transistors, respectively. They are connected to an XA-YA group ofmA numbers of power paths and an XB-YB group of mB numbers of powerpaths respectively. The switch device further comprises nA numbers oftransformers 3A and nB numbers of transformers 3B, and also a pluralityof power sources 100A and 100B, and loads 200A and 200B. FIG. 16 is aconceptual diagram showing one group of closed loops comprising mainpower transistors, power sources and loads. In this arrangement, thepower sources and loads are commonly operated in the individual powernetworks.

The two power networks A and B may be consolidated together into one asshown by simplified diagrams in FIG. 1.

In FIG. 15, assume that the two circuits (a) and (b) are a powerclosed-loop. Then it becomes possible for the two power loops to supplythe base drive current (drive voltage) of the main power transistor onthe side opposite to each other. A simple way to effect this operationis to turn on and off the main power transistors 1A and 1B synchronouslywith each other as in FIG. 16.

Assume that both groups A and B comprise a plurality of closed-loops inwhich at least one of the transformers is in operation and at least oneof the transformer secondary windings connected to one power transistoris in operation. Then this power transistor can be arbitrarily turnedon-off and the supply of its base-drive power is secured. In this case,it is not necessary to satisfy the foregoing synchronized turn-on-offcondition for the power transistors.

What is important is that at least one of the transformers connected tothe main transistor be operated at least for the period when this maintransistor is turned on. To meet this requirement, there may be avariety of types of power networks and transformer networks available.

Basic principles and ideas of the invention have been described above.For a better understanding of the invention, several applicationexamples will be described below.

Referring to FIGS. 18(a) and 18(b), there are shown examples ofapplication of the invention to an inverter of the single-way connectiontype; (a) is a current transformer system, and (b) a voltage transformersystem, wherein an output transformer 200 and a voltage transformer 3are used on common. The voltage transformer 3 has its core used incommon for main power transistors 1a and 1b. In the voltage transformersystem as in FIG. 18(b), a base-drive current limiting resistor r (witha small resistance) may be used when the tap voltage is too high at thesecondary windings 32νa and 32νb. The circuit as in FIG. 17 may beapplied to various single way connections and multiphase connections.

The circuits shown in FIGS. 18(a) and 18(b) constitute a series-pairconnection (half-bridge) used for bidirectional power conversion controlor bidirectional conductive power conversion control and are suited forapplications to inverters and DC power controls.

The circuit shown in FIG. 18(a) is an example wherein an AC currenttransformer is employed with its core used in common. This circuit canbe used in the voltage transformer system when the number of turns ofthe primary winding 31 is increased and the primary winding is connectedin parallel to the load 200.

The circuit in FIG. 18(b) is an example wherein a current transformer isdisposed for each main power transistor. This example is suited forapplications to control of a DC load (reversible-polar load 200 orunipolar voltage load 200'), time-ratio modulation type invertersoperable in a wide frequency range, etc.

In FIG. 18, the rectifier element 201 may be connected in the positionindicated by the dotted line 201'. The circuit in FIG. 18 can be appliedto various bridge circuitry.

FIG. 19 is a circuit diagram showing a switch circuit embodying theconcept of the invention illustrated in FIG. 16. In FIG. 19, the numeral300 denotes a one unit arm comprising a main transistor 1, a rectifierelement 201, an auxiliary solid-state switch 2, and a branch means 4.There are also provided a main current terminal X-Y, and base-drivecurrent supply terminals C₁ and C₂. The reference 20 indicates arectifier for a base-drive current supply, and 400A and 400B and bridgetype inverter units having a construction similar to each other. Thegroup A inverter unit 400A receives power from a primary currenttransformer 3B installed in the output current path of the group Binverter unit 400B. This power supply may be generated from a currenttransformer 3A inserted in its own output current path. The secondaryoutput current of the primary current transformer 3B is suppliedcommonly through a rectifier 20a to three unit arms of the positivegroup. The secondary output current of this primary current transformer3B is connected to the primary windings of the secondary currenttransformers 3a, 3b and 3c. The secondary outputs of these secondarycurrent transformers are connected to rectifiers 20b, 20c and 20d andare supplied to three unit arms, respectively, of the negative group.

In this case, it is necessary to operate the two groups 400A and 400Bsimultaneously where the turn-on-off synchronism among power transistorsneed not be established. This example is advantageous in that the loadcurrent is balanced between the two groups A and B.

In this example, a DC voltage transformer or, in particular, a DCcurrent transformer, may be used instead of the above voltagetransformer. Also, a DC current transformer operable without generatingripples may be used. Further, a current transformer having a response toa wide frequency range may also be used. With the use of a DC currenttransformer, the limitation on the turn-on holding time is eliminated.

According to the invention, as has been described with reference to itsembodiments illustrated in FIGS. 6 to 19 in connection with the switchdevice wherein an auxiliary solid-state switch is connected between thecollector and the base of a main power transistor whereby the base-drivecurrent is supplied to the main power transistor, the voltage or currentin the main power circuit comprising the main power transistor and theother main power transistors is transformed and supplied to theauxiliary solid-state switch branches and thus the voltage drop in themain power transistor in the ON state is reduced, the loss in the mainpower transistor is diminished, and the base-drive power of the mainpower transistor is self-sufficiently replenished.

Obviously, numerous modifications and variations of the presentinvention are possible in light of the above teachings. It is thereforeto be understood that within the scope of the appended claims theinvention may be practiced otherwise than as specifically describedherein.

What is claimed is:
 1. A transistor switch device comprising:a mainpower transistor having a collector, a base and an emitter with avoltage drop between the collector and the emitter, an auxiliarysolid-state switch, a series connected branch comprising said auxiliarysolid-state switch and an electricity replenishing means, said seriesconnected branch being connected between the collector and the base ofthe main power transistor, and said electricity replenishing meansgenerating a voltage, and shunt means comprising a diode connected inparallel with said electricity replenishing means for by-passing acurrent of said electricity replenishing means when said auxiliary solidstate switch is turned off, whereby the voltage drop between thecollector and the emitter of the main power transistor is reduced.
 2. Atransistor switch device according to claim 1 wherein the electricityreplenishing means is a DC power source.
 3. A transistor switch deviceaccording to claim 1 wherein the electricity replenishing means furthercomprises a rectifier having input and output terminals, an AC powersource and an AC transformer having a plurality of primary windings anda plurality of secondary windings,the output terminals of the rectifierbeing serially connected across the series connected branch, the inputterminals of the rectifier being connected to at least one of thesecondary windings of the AC transformer, the AC power source beingconnected to at least one of the primary windings of the AC transformer.4. A transformer switch device according to claim 1 wherein theelectricity replenishing means comprises a high-frequency AC powersource whose output is connected to the input of a high-frequencytransformer whose output is connected to the input of a rectifier,thefrequency of the high-frequency AC power source being higher than 100Herz.
 5. A transistor switch device according to claim 1 wherein theauxiliary solid-state switch is an auxiliary transistor.
 6. A transistorswitch device comprising:a main power transistor having a collector, abase and an emitter with a voltage drop between the collector and theemitter, an auxiliary solid-state switch, a series connected branchcomprising said auxiliary solid state switch and an electricityreplenishing means, wherein the electricity replenishing means comprisesa transformer capable of deriving electricity from a power circuit inwhich the main power transistor is disposed, and the transformer is acurrent transformer having a primary winding connected to the powercircuit in which the main power transistor is disposed and a secondarywinding connected in said series connected branch, said series connectedbranch being connected between the collector and the base of the mainpower transistor, and said electricity replenishing means generating avoltage, whereby the voltage drop between the collector and the emitterof the main power transistor is reduced.
 7. A transistor switch devicecomprising:a main power transistor having a collector, a base and anemitter with a voltage drop between the collector and the emitter, anauxiliary solid-state switch, a series connected branch comprising saidauxiliary solid-state switch and an electricity replenishing means,wherein the electricity replenishing means comprises a transformercapable of deriving electricity from a power circuit in which the mainpower transistor is disposed, shunt means connected in parallel to thesecondary winding of the transformer for shunting the secondary currentof the transformer when the auxiliary solid-state switch is turned off,said series connected branch being connected between the collector andthe base of the main power transistor, and said electricity replenishingmeans generating a voltage, whereby the voltage drop between thecollector and the emitter of the main power transistor is reduced.
 8. Atransistor switch device comprising:a main power transistor having acollector, a base and an emitter with a voltage drop between thecollector and the emitter, an auxilary solid-state switch, a seriesconnected branch comprising said auxiliary solid-state switch and anelectricity replenishing means, wherein the electricity replenishingmeans comprises a transformer capable of deriving electricity from apower circuit in which the main power transistor is disposed, shuntmeans connected between the base and the emitter of the main powertransistor to bypass excitation current of the transformer when theauxiliary solid-state switch is turned on, said series connected branchbeing connected between the collector and the base of the main powertransistor, and said electricity replenishing means generating avoltage, whereby the voltage drop between the collector and the emitterof the main power transistor is reduced.
 9. A transistor switch deviceaccording to claim 8 wherein the shunt means is a tertiary winding ofthe transformer.
 10. A transistor switch device comprising:a main powertransistor having a collector, a base and an emitter with a voltage dropbetween the collector and the emitter, an auxiliary solid-state switch,a series connected branch comprising said auxiliary solid-state switchand an electricity replenishing means, wherein the electricityreplenishing means comprises a transformer capable of derivingelectricity from a power circuit in which the main power transistor isdisposed, and the transformer is a voltage transformer which derivesvoltage from the power circuit in which the main power transistor isdisposed, the voltage transformer having a primary winding connected tothe power circuit and a secondary winding connected in said seriesconnected branch, said series connected branch being connected betweenthe collector and the base of the main power transistor, and saidelectricity replenishing means generating a voltage, whereby the voltagedrop between the collector and the emitter of the main power transistoris reduced.